Direct
Sequence Spread Spectrum:
Aim:
Introduction
to Direct sequence spread spectrum.
Introduction:
In
some situations it is required that a communication signal be
difficult to detect, and difficult to demodulate even when detected.
Here the word ‘detect’ is used in the sense of ‘to discover the
presence of’. The signal is required to have a low probability of
intercept - LPI.
In
other situations a signal is required that is difficult to interfere
with, or ‘jam’. The ‘spread spectrum’ signal has properties
which help to achieve these ends. Spread spectrum signals may be
divided into two main groups - direct sequence spread spectrum
(DSSS), and frequency hopping spread spectrum (FHSS). This experiment
is concerned with demonstrating some of the principles of the first.
Principle
of DSSS:
Consider
the frequency translation of a baseband message (of bandwidth B Hz)
to a higher part of the spectrum, using DSBSC modulation. The
resulting signal occupies a bandwidth of 2B Hz, and would typically
override the noise occupying the same part of the spectrum. This
makes it easy to find with a spectrum analyser (for example), and so
the probability of intercept is high. A local carrier, synchronized
with
that at the transmitter, is required at the receiver for synchronous
demodulation. The recovered signal-to-noise ratio is 3 dB better than
that measured at its original location in the spectrum. This 3 dB
improvement comes from the fact that the contributions from each
sideband add coherently, whereas the noise does not. This can be
called a 3 dB ‘processing gain’, and is related to the fact that
the transmission
bandwidth
and message bandwidth are in the ratio of 2:1.
In
a spread spectrum system literally thousands of different carriers
are used, to generate thousands of DSBSC signals each derived from
the same message. These carriers are spread over a wide bandwidth
(much wider than 2B Hz), and so the resulting DSBSC signals will be
spread over the same bandwidth.
If
the total transmitted power is similar to that of the single DSBSC
case, then the power of an individual DSBSC in the spread spectrum
case is thousands of times less. In fact, over the bandwidth occupied
by one of these DSBSC signals, it would be literally ‘buried in the
noise’, and difficult to find with a spectrum analyser (for
example).
Instead
of the total transmitted power being concentrated in a band of width
2B Hz, the multiple carriers have spread it thinly over a very wide
bandwidth. The signal-to-noise ratio for each DSBSC is very low (well
below 0 dB). To recover the message from the
transmitted
spread spectrum signal all that a receiver requires is thousands of
local carriers, at the same frequency and of the same relative phase,
as all those at the transmitter . All these carriers come from a
pseudo random binary sequence (PRBS) generator.
Given
a stable clock, and a long sequence, it may be shown that the
spectrum of a pseudo random binary sequence generator is a good
source of these carriers . A second PRBS generator, of the same type,
clocked at the same rate, and appropriately aligned, is sufficient to
regenerate all the required local carriers at the receiver
demodulator.
In
the spread spectrum context the PRBS signal is generally called a PN
– pseudo noise - signal, since its spectrum approaches that of
random noise.
Having
the correct sequence at the receiver means that the message
contributions from each of the thousands of minute DSBSC signals
combine in phase – coherently - and add up to a finite message
output. Otherwise they add with random phases, resulting in a (very)
small, noise-like output.
Processing
gain:
To
achieve most of the claims made for the spread spectrum it is
necessary that the bandwidth over which the message is spread be very
much greater than the bandwidth of the message itself. Each DSBSC of
the DSSS signal is at a level below the noise, but each is processed
by the synchronous demodulator to give a 3 dB SNR improvement. The
total improvement is proportional to the number of individual DSBSC
components. In fact the processing gain of the system is equal to the
ratio of DSSS bandwidth to message bandwidth.
A
DSSS generator:
To
generate a spread spectrum signal one requires:
1.
A modulated signal somewhere in the RF spectrum
2.
A PN sequence to spread it
These
two are combined as shown in Figure 1.
There
are two bandwidths involved here: that of the modulated signal, and
the spreading sequence. The first will be very much less than the
second. The output spread spectrum signal will be spread either side
of the original RF carrier (ω0) by an amount equal to the bandwidth
of the PN sequence.
Most
of the energy of the sequence will lie in the range DC to ωs, where
ωs is the sequence clock. The longer the sequence the more spectral
components will lie in this range. It is necessary and usual that ω0
>> ωs, although in the experiment to follow the difference
will not be large.
The
modulated signal can be of any type, but typically digitally-derived,
such as binary phase shift keyed - BPSK. In this case the arrangement
of Figure 1 can be expanded to that of Figure 2.
A
digital message is preferred in an operational spread spectrum
system, since it makes the task of the eavesdropper even more
difficult.
The
arrangement of Figure 2 can be simplified by noting that, if the
clock of the bipolar message is a sub-multiple of the clock of the PN
sequence, then the modulotwo sum of the message and the PN sequence
can be used to multiply the RF carrier, generating a DSSS signal with
a single multiplier. Such a simplification will not be implemented in
this experiment.
A
DSSS demodulator:
A
demodulator for the DSSS of Figure 1 is shown in block form in Figure
3.
The
input multiplier performs the de-spreading of the received signal,
and the second multiplier translates the modulated signal down to
baseband. The filter output would probably require further processing
- not shown - to ‘clean up’ the waveform to binary format.
The
PN sequence at the receiver acts as a ‘key’ to the transmission.
It must not only have the same clock and bit pattern; it must be
aligned properly with the sequence at the transmitter.
The
PN spectrum:
The
PN signal, being periodic, has a line spectrum. This spectrum is
determined by the PN clock period Tc and the sequence length N (the
number of bits, or clock periods, before the pattern repeats).
• the
spectral lines are separated by (1/NTc) Hz.
• there
is a DC component of amplitude (1/N).
the
amplitude of an individual line in the spectrum is weighted, where:
It
is clear that a plot of these weights will show them lying within an
envelope having a sync function shape. Most of the energy of the PN
sequence lies below the first minimum (when n = N); that is, below
the clock frequency.
For
approximate analysis it is often assumed that the shape of the power
spectral density is rectangular, extending from DC to (1/Tc) Hz.
Experimental
Procedure:
This
experiment will be concerned with modelling the systems of Figures 2
and 3.
The
message:
The
message comes from a SEQUENCE GENERATOR.
To
obtain a reasonable processing gain the message clock needs to be
much slower than the PN clock. Being a sub-multiple of the PN clock
is also an advantage. The2 kHz MESSAGE from the MASTER SIGNALS module
has been used - 1/48 of the 100 kHz master clock.. You may prefer a
larger division ratio. This can be achieved with further division
using the DIGITAL UTILITIES module.
Select
a short message sequence for stable oscilloscope displays (both
toggles of the on-board switch UP).
The
transmission medium:
The
transmitter is connected to the receiver via an ADDER, acting as a
nonbandlimiting (and so no delay) transmission path. The second input
to the ADDER will be used for inserting noise. The inclusion of a
finite delay would introduce problems with aligning the receiver PN
sequence.
Clocks:
Since
the PN clock is a sub-multiple of the carrier, only one of these
needs to be recovered by the receiver. In the experiment they are
stolen from the transmitter.
Generation:
T1
model the block diagram of Figure 2. This is shown in Figure 4. The
ADDER is included for inserting noise from a NOISE GENERATOR (module
not shown).
T2
before inserting the SEQUENCE GENERATOR modules, select a short
sequence for the message (both toggles of the on-board switch SW2
UP), and the same long sequence for the PN generators (both toggles
of the on-board switch SW2 DOWN).
T3
initially use the 100 kHz TTL available from MASTER SIGNALS, divided
by 12, using a DIGITAL UTILITIES module (not shown), for the PN
generator clock.
T4
initially reduce the noise output from the ADDER to zero.
T5
instead of connecting the bi-polar message sequence to the X input of
the first MULTIPLIER, connect instead the 2 kHz MESSAGE (sinusoidal)
signal. This makes the output from the first MULTIPLIER a DSBSC
signal, easily recognisable on the oscilloscope. Check this.
T6
instead of connecting the PN sequence to the X input of the second
MULTIPLIER, connect instead the VARIABLE DC module set to near +2
volt. This makes the second MULTIPLIER a voltage controlled amplifier
with a gain of about unity. Thus the ‘DSSS output’ will be a
well-recognisable DSBSC based on a 2 kHz message. Check your levels
with this recognisable signal.
T7
using the SPECTRUM ANALYSER, examine the output spectrum. Confirm it
is a DSBSC.
When
satisfied that the MULTIPLIER modules are behaving as expected,
return their inputs to the signals previously connected.
T8
synchronize the oscilloscope to the SYNCH signal (START-OF-SEQUENCE)
of the message generator. Examine signals throughout the system. Some
will be familiar, others not. There are no adjustments to be made,
except for the output amplitude from the ADDER.
T9
using the SPECTRUM ANALYSER, examine the output spectrum. With an
8.333 kHz PN clock, confirm that the output spectrum - the DSSS
signal - has its energy concentrated over about 8 kHz either side of
the 100 kHz carrier.
T10
now add noise. Adjust the noise level so that, while observing the
spectrum of the ADDER output, the DSSS signal can be seen above the
noise level.
T11
while still observing the spectrum, increase the spread of the DSSS
signal. This is done by increasing the PN sequence clock rate by
choosing a lower division of the 100 kHz TTL - choose divide-by-2,
for a 50 kHz clock.
The
increase of PN clock rate has widened the spectrum of the PN sequence
to about 50 kHz (from 8 kHz). Since the DSSS signal contains the same
energy as before, it has been spread more thinly over the spectrum,
and it will have sunk deeper into, and got ‘lost’ in, the noise.
Demodulation:
T12
model the receiver of Figure 3 as suggested in Figure 5 below. Both
the 100 kHz carrier, and the PN sequence, are stolen from the
transmitter. Not shown is a PHASE SHIFTER for the 100 kHz carrier.
This is used to maximize the output amplitude (it will also change
its polarity).
T13
the bandwidth of the output filter is chosen to suit the message. Use
a TUNEABLE LPF (shown in Figure 5), or the 3 kHz LPF in the HEADPHONE
AMPLIFIER. For restoration of the output to a TTL format a DECISION
MAKER would be included, but this is not necessary for this
experiment. Visual comparison of the sent and received sequences is
adequate.
Although
there are two stolen clocks shown, in practice it is often only
necessary to acquire, by what ever means, a single clock. This is
because one can be a known sub-multiple of the other.
T14
observe the output, when the transmitter is connected to the input.
Probably there will be ‘nothing’ - or nothing resembling the
expected output sequence. Varying the phase of the 100 kHz carrier
should not change things.
The
problem is that the receiver PN sequence, although synchronized with
that at the transmitter, is not correctly aligned in time. With no
transmission delay it is a simple matter to achieve this.
T15
bring the two sequences into alignment by momentarily connecting the
startof- sequence SYNC output of the transmitter SEQUENCE GENERATOR
to the RESET input of the receiver SEQUENCE GENERATOR.
T16
re-examine the output from the demodulator. The message should have
been recovered (being a short sequence, this is easy to confirm
visually). Adjust the bandwidth of the demodulator output filter for
minimum bandwidth consistent with reasonable waveshape. Remember, a
DECISION MAKER could be used to regenerate a perfect copy of the
original, but this is not necessary for our present purpose.
Interference:
T17
with the system set up and showing the demodulated sequence at the
receiver output, replace the noise with a 100 kHz sinusoid from a
VCO. This represents an interfering signal (a very elementary form of
jamming). Monitor the VCO with the FREQUENCY COUNTER.
T18
while watching the demodulator output, sweep the VCO frequency
through its full frequency range.
Discussion
Question:
1)
Consider a DSBSC signal derived from a single tone. How many lines
would there be in the spectrum of the spread signal? You will have to
supply some data regarding the spreading sequence.
2)
Consider a DSBSC signal derived from a single tone. How many lines
would there be in the de-spread spectrum ? You will have to supply
some data regarding the spreading sequence.
3)
What advantage is there in making the message bit rate a sub-multiple
of the PN bit rate ?
4)
Explain the principle of operation involved in CDMA?
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